= + j t (18-1) X ( ) x t e ( ) dt , X then - - PowerPoint PPT Presentation

1

For a deterministic signal x(t), the spectrum is well defined: If represents its Fourier transform, i.e., if then represents its energy spectrum. This follows from Parseval’s theorem since the signal energy is given by Thus represents the signal energy in the band (see Fig 18.1). ( ) X ω ( ) ( ) ,

j t

X x t e dt

ω

ω

+∞ − −∞

= ∫

2

| ( ) | X ω

2 2

1 2

( ) | ( ) | . x t dt X d E

π

ω ω

+∞ +∞ −∞ −∞

= =

∫ ∫

(18-1) (18-2)

2

| ( ) | X ω ω ∆ ( , ) ω ω ω + ∆ Fig 18.1

• 18. Power Spectrum

t

( ) X t

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ω ω

2

| ( )| X ω

Energy in

( , ) ω ω ω +∆ ω ω + ∆

2

However for stochastic processes, a direct application of (18-1) generates a sequence of random variables for every Moreover, for a stochastic process, E{| X(t) |2} represents the ensemble average power (instantaneous energy) at the instant t. To obtain the spectral distribution of power versus frequency for stochastic processes, it is best to avoid infinite intervals to begin with, and start with a finite interval (– T, T ) in (18-1). Formally, partial Fourier transform of a process X(t) based on (– T, T ) is given by so that represents the power distribution associated with that realization based

• n (– T, T ). Notice that (18-4) represents a random variable for every

and its ensemble average gives, the average power distribution based on (– T, T ). Thus . ω ( ) ( )

T j t T T

X X t e dt

ω

ω

− −

= ∫

2 2

| ( ) | 1 ( ) 2 2

T j t T T

X X t e dt T T

ω

ω

− −

=

∫

(18-3) (18-4) , ω

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3

represents the power distribution of X(t) based on (– T, T ). For wide sense stationary (w.s.s) processes, it is possible to further simplify (18-5). Thus if X(t) is assumed to be w.s.s, then and (18-5) simplifies to Let and proceeding as in (14-24), we get to be the power distribution of the w.s.s. process X(t) based on (– T, T ). Finally letting in (18-6), we obtain T → ∞

1 2

( ) 1 2 1 2

1 ( ) ( ) . 2

T XX

T T j t t T T

P R t t e dt dt T

ω

ω

− − − −

= −

∫ ∫

1 2 1 2

2 ( ) * 1 2 1 2 ( ) 1 2 1 2

| ( ) | 1 ( ) { ( ) ( )} 2 2 1 ( , ) 2

T XX

T T j t t T T T T T j t t T T

X P E E X t X t e dt dt T T R t t e dt dt T

ω ω

ω ω

− − − − − − − −

  = =     =

∫ ∫ ∫ ∫

2 2 2 | | 2 2

1 ( ) ( ) (2 | |) 2 ( ) (1 )

T XX XX

T j T T j T T

P R e T d T R e d

ωτ τ ωτ

ω τ τ τ τ τ

− − − −

= − = − ≥

∫ ∫

1 2 1 2

( , ) ( )

XX XX

R t t R t t = −

1 2

t t τ = − (18-5) (18-6)

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to be the power spectral density of the w.s.s process X(t). Notice that i.e., the autocorrelation function and the power spectrum of a w.s.s Process form a Fourier transform pair, a relation known as the Wiener-Khinchin Theorem. From (18-8), the inverse formula gives and in particular for we get From (18-10), the area under represents the total power of the process X(t), and hence truly represents the power

• spectrum. (Fig 18.2).

( ) lim ( ) ( )

XX T XX

j T

S P R e d

ωτ

ω ω τ τ

+∞ − −∞ →∞

= = ≥

∫

F T

( ) ( ) 0.

XX XX

R S ω ω

⋅

←→ ≥

1 2

( ) ( )

XX XX

j

R S e d

ωτ

π

τ ω ω

+∞ −∞

=

∫

2

1 2

( ) (0) {| ( ) | } ,

XX XX

S d R E X t P the total power.

π

ω ω

+∞ −∞

= = =

∫

0, τ = ( )

XX

S ω ( )

XX

S ω (18-9) (18-8) (18-7) (18-10)

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Fig 18.2 The nonnegative-definiteness property of the autocorrelation function in (14-8) translates into the “nonnegative” property for its Fourier transform (power spectrum), since from (14-8) and (18-9) From (18-11), it follows that ( ) ( ) 0.

XX XX

R nonnegative - definite S τ ω ⇔ ≥

( ) * * 1 1 1 1 2 1

1 2 1 2

( ) ( ) ( ) 0.

i j XX XX i XX

n n n n j t t i j i j i j i j i j n j t i i

a a R t t a a S e d S a e d

ω ω

π π

ω ω ω ω

+∞ − −∞ = = = = +∞ = −∞

− = = ≥

∑∑ ∑∑ ∫ ∑ ∫

(18-11) (18-12)

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ω ω + ∆ ω represents the power in the band ( , ) ω ω ω + ∆ ( )

XX

S ω ω ∆

( )

XX

S ω

ω

6

If X(t) is a real w.s.s process, then so that so that the power spectrum is an even function, (in addition to being real and nonnegative). ( ) = ( )

XX XX

R R τ τ − ( ) ( ) ( )cos 2 ( )cos ( )

XX XX XX XX XX

j

S R e d R d R d S

ωτ

ω τ τ τ ωτ τ τ ωτ τ ω

+∞ − −∞ +∞ −∞ ∞

= = = = − ≥

∫ ∫ ∫

(18-13)

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Power Spectra and Linear Systems

If a w.s.s process X(t) with autocorrelation function is applied to a linear system with impulse response h(t), then the cross correlation function and the output autocorrelation function are given by (14-40)-(14-41). From there But if Then since ( )

XY

R τ ( )

YY

R τ ( ) ( )

XX XX

R S τ τ ↔ ≥ h(t) X(t) Y(t) Fig 18.3

* *

( ) ( ) ( ), ( ) ( ) ( ) ( ).

XY XX YY XX

R R h R R h h τ τ τ τ τ τ τ = ∗ − = ∗ − ∗ (18-14) ( ) ( ), ( ) ( ) f t F g t G ω ω ↔ ↔ (18-16) (18-15) ( ) ( ) ( ) ( ) f t g t F G ω ω ∗ ↔ { ( ) ( )} ( ) ( )

j t

f t g t f t g t e dt

ω +∞ − −∞

∗ = ∗

∫

F

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(18-20) (18-19) (18-18) Using (18-15)-(18-17) in (18-14) we get since where represents the transfer function of the system, and

{ }

( )

{ ( ) ( )}= ( ) ( ) = ( ) ( ) ( ) = ( ) ( ).

j t j j t

f t g t f g t d e dt f e d g t e d t F G

ω ωτ ω τ

τ τ τ τ τ τ τ ω ω

+∞ +∞ − −∞ −∞ +∞ +∞ − − − −∞ −∞

∗ − − −

∫ ∫ ∫ ∫

F (18-17)

* *

( ) { ( ) ( )} ( ) ( )

XY XX XX

S R h S H ω ω τ ω ω = ∗ − = F

( )

* * *

( ) ( ) ( ),

j j t

h e d h t e dt H

ωτ ω

τ τ ω

+∞ +∞ − − −∞ −∞

− = =

∫ ∫

( ) ( )

j t

H h t e dt

ω

ω

+∞ − −∞

= ∫

2

( ) { ( )} ( ) ( ) ( ) | ( ) | .

YY YY XY XX

S R S H S H ω τ ω ω ω ω = = = F

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From (18-18), the cross spectrum need not be real or nonnegative; However the output power spectrum is real and nonnegative and is related to the input spectrum and the system transfer function as in (18-20). Eq. (18-20) can be used for system identification as well. W.S.S White Noise Process: If W(t) is a w.s.s white noise process, then from (14-43) Thus the spectrum of a white noise process is flat, thus justifying its

• name. Notice that a white noise process is unrealizable since its total

power is indeterminate. From (18-20), if the input to an unknown system in Fig 18.3 is a white noise process, then the output spectrum is given by Notice that the output spectrum captures the system transfer function characteristics entirely, and for rational systems Eq (18-22) may be used to determine the pole/zero locations of the underlying system. ( ) ( ) ( ) .

WW WW

R q S q τ δ τ ω = ⇒ = (18-22) (18-21)

2

( ) | ( ) |

YY

S q H ω ω =

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Example 18.1: A w.s.s white noise process W(t) is passed through a low pass filter (LPF) with bandwidth B/2. Find the autocorrelation function of the output process. Solution: Let X(t) represent the output of the LPF. Then from (18-22) Inverse transform of gives the output autocorrelation function to be

2

, | | / 2 ( ) | ( ) | . 0, | | / 2

XX

q B S q H B ω ω ω ω ≤  = =  >  ( )

XX

S ω

/ 2 / 2 / 2 / 2

( ) ( ) sin( / 2) sinc( / 2) ( / 2)

XX XX

B B j j B B

R S e d q e d B qB qB B B

ωτ ωτ

τ ω ω ω τ τ τ

− −

= = ∫ ∫ = = (18-23) (18-24)

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• Fig. 18.4

(a) LPF

ω 2 | ( )| H ω / 2 B − / 2 B 1 τ qB ( )

XX

R τ

(b)

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Eq (18-23) represents colored noise spectrum and (18-24) its autocorrelation function (see Fig 18.4). Example 18.2: Let represent a “smoothing” operation using a moving window on the input process X(t). Find the spectrum of the output Y(t) in term of that of X(t). Solution: If we define an LTI system with impulse response h(t) as in Fig 18.5, then in term of h(t), Eq (18-25) reduces to so that Here

1 2

( ) ( )

t T t T

T

Y t X d τ τ

+ −

=

∫

(18-25) ( ) ( ) ( ) ( ) ( ) Y t h t X d h t X t τ τ τ

+∞ −∞

= − = ∗

∫

2

( ) ( ) | ( ) | .

YY XX

S S H ω ω ω =

1 2

( ) sinc( )

T j t T T

H e dt T

ω

ω ω

+ − −

= =

∫

(18-28) (18-27) (18-26)

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Fig 18.5

T − T

t

( ) h t

1 / 2T

12

so that

2

( ) ( ) sinc ( ).

YY XX

S S T ω ω ω = (18-29) Notice that the effect of the smoothing operation in (18-25) is to suppress the high frequency components in the input and the equivalent linear system acts as a low-pass filter (continuous- time moving average) with bandwidth in this case.

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(beyond / ), T π Fig 18.6

ω ( )

XX

S ω

2

sinc ( ) T ω T π ω ω ( )

YY

S ω

2 /T π

13

Discrete – Time Processes For discrete-time w.s.s stochastic processes X(nT) with autocorrelation sequence (proceeding as above) or formally defining a continuous time process we get the corresponding autocorrelation function to be Its Fourier transform is given by and it defines the power spectrum of the discrete-time process X(nT). From (18-30), so that is a periodic function with period { } ,

k

r

+∞ −∞

( ) ( ) ( ),

n

X t X nT t nT δ = −

∑

( ) ( ).

XX

k k

R r kT τ δ τ

+∞ =−∞

= −

∑

(18-30) ( ) 0,

XX

j T k k

S r e

ω

ω

+∞ − =−∞

= ≥

∑

( )

XX

S ω ( ) ( 2 / )

XX XX

S S T ω ω π = + 2 2 . B T π = (18-31) (18-32)

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This gives the inverse relation and represents the total power of the discrete-time process X(nT). The input-output relations for discrete-time system h(nT) in (14-65)-(14-67) translate into and where represents the discrete-time system transfer function. 1 ( ) 2

XX

B jk T k B

r S e d B

ω

ω ω

−

=

∫

(18-33)

2

1 {| ( ) | } ( ) 2

XX

B B

r E X nT S d B ω ω

−

= =

∫

(18-34)

*

( ) ( ) ( )

XY XX

j

S S H e ω ω ω =

2

( ) ( ) | ( ) |

YY XX

j

S S H e ω ω ω = ( ) ( )

j j nT n

H e h nT e

ω ω +∞ − =−∞

= ∑ (18-35) (18-37) (18-36)

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Matched Filter

Let r(t) represent a deterministic signal s(t) corrupted by noise. Thus where r(t) represents the observed data, and it is passed through a receiver with impulse response h(t). The

• utput y(t) is given by

where and it can be used to make a decision about the presence of absence

• f s(t) in r(t). Towards this, one approach is to require that the

receiver output signal to noise ratio (SNR)0 at time instant t0 be

• maximized. Notice that

h(t)

r(t) y(t)

Fig 18.7 Matched Filter

t t =

( ) ( ) ( ), 0 r t s t w t t t = + < < (18-38) (18-39) ( ) ( ) ( ), ( ) ( ) ( ),

s

y t s t h t n t w t h t = ∗ = ∗ (18-40)

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( ) ( ) ( )

s

y t y t n t = +

∆

16

represents the output SNR, where we have made use of (18-20) to determine the average output noise power, and the problem is to maximize (SNR)0 by optimally choosing the receiver filter Optimum Receiver for White Noise Input: The simplest input noise model assumes w(t) to be white noise in (18-38) with spectral density N0, so that (18-41) simplifies to and a direct application of Cauchy-Schwarz’ inequality in (18-42) gives (18-41) ( ). H ω

2 2

( ) ( ) ( ) 2 | ( ) |

j t

S H e d SNR N H d

ω

ω ω ω π ω ω

+∞ −∞ +∞ −∞

= ∫

∫

(18-42)

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2 2 2 2 2

1 2 1 1 2 2

Output signal power at | ( ) | ( ) Average output noise power {| ( ) | } ( ) ( ) | ( ) | ( ) ( ) | ( ) |

nn WW

s j t s

t t y t SNR E n t S H e d y t S d S H d

ω

π π π

ω ω ω ω ω ω ω ω

+∞ −∞ +∞ +∞ −∞ −∞

= = = = =

∫ ∫ ∫

∆

17

and equality in (18-43) is guaranteed if and only if

• r

From (18-45), the optimum receiver that maximizes the output SNR at t = t0 is given by (18-44)-(18-45). Notice that (18-45) need not be causal, and the corresponding SNR is given by (18-43).

2 2

1 2

( ) ( ) | ( ) |

s

N

s t dt E SNR S d N N

π

ω ω

+∞ +∞ −∞

≤ = =

∫ ∫

(18-43)

*

( ) ( )

j t

H S e

ω

ω ω

−

= ( ) ( ). h t s t t = − (18-44) (18-45)

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Fig 18.8 (a) (b) t0=T/2 (c) t0=T

( ) h t ( ) s t ( ) h t T t t t T / 2 T −

t0

Fig 18-8 shows the optimum h(t) for two different values of t0. In Fig 18.8 (b), the receiver is noncausal, whereas in Fig 18-8 (c) the receiver represents a causal waveform.

18

If the receiver is not causal, the optimum causal receiver can be shown to be and the corresponding maximum (SNR)0 in that case is given by Optimum Transmit Signal: In practice, the signal s(t) in (18-38) may be the output of a target that has been illuminated by a transmit signal f (t) of finite duration T. In that case where q(t) represents the target impulse response. One interesting question in this context is to determine the optimum transmit (18-48) ( ) ( ) ( ) ( ) ( ) ,

T

s t f t q t f q t d τ τ τ = ∗ = −

∫

(18-47) (18-46) ( ) ( ) ( )

• pt

h t s t t u t = −

2

1

( ) ( )

t

N

SNR s t dt =

∫

( ) f t T t

q(t) ( ) f t ( ) s t Fig 18.9

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signal f (t) with normalized energy that maximizes the receiver output SNR at t = t0 in Fig 18.7. Notice that for a given s(t), Eq (18-45) represents the optimum receiver, and (18-43) gives the corresponding maximum (SNR)0. To maximize (SNR)0 in (18-43), we may substitute (18-48) into (18-43). This gives where is given by and is the largest eigenvalue of the integral equation

1 2

2 1 1 1 * 1 2 2 2 1 1 ( , ) 1 2 2 2 1 1 max

1 1

( ) |{ ( ) ( ) }| ( ) ( ) ( ) ( ) { ( , ) ( ) } ( ) /

T T T T T

N N

SNR q t f d dt q t q t dt f d f d f d f d N

τ τ

τ τ τ τ τ τ τ τ τ τ τ τ τ τ τ λ

∞ ∞ Ω

= − = − − = Ω ≤

∫ ∫ ∫ ∫ ∫ ∫ ∫

• max

λ

1 2

( , ) τ τ Ω

* 1 2 1 2

( , ) ( ) ( ) q t q t dt τ τ τ τ

∞

Ω = − −

∫

(18-49) (18-50)

1 2 2 2 max 1 1

( , ) ( ) ( ), 0 .

T

f d f T τ τ τ τ λ τ τ Ω = < <

∫

(18-51)

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If the causal solution in (18-46)-(18-47) is chosen, in that case the kernel in (18-50) simplifies to and the optimum transmit signal is given by (18-51). Notice that in the causal case, information beyond t = t0 is not used. and Observe that the kernal in (18-50) captures the target characteristics so as to maximize the output SNR at the observation instant, and the optimum transmit signal is the solution of the integral equation in (18-51) subject to the energy constraint in (18-52). Fig 18.10 show the optimum transmit signal and the companion receiver pair for a specific target with impulse response q(t) as shown there .

2

( ) 1.

T f

t dt =

∫

(18-52)

1 2

( , ) τ τ Ω

* 1 2 1 2

( , ) ( ) ( ) .

t q t

q t dt τ τ τ τ Ω = − −

∫

(18-53)

t

(a)

( ) q t

(b)

t T ( ) f t t ( ) h t t

(c)

Fig 18.10

21

What if the additive noise in (18-38) is not white? Let represent a (non-flat) power spectral density. In that case, what is the optimum matched filter? If the noise is not white, one approach is to whiten the input noise first by passing it through a whitening filter, and then proceed with the whitened output as before (Fig 18.7). Notice that the signal part of the whitened output sg(t) equals where g(t) represents the whitening filter, and the output noise n(t) is white with unit spectral density. This interesting idea due to ( )

WW

S ω ( ) ( ) ( )

g

s t s t g t = ∗ (18-54)

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Whitening Filter g(t) ( ) ( ) ( ) r t s t w t = + ( ) ( )

g

s t n t + Fig 18.11

colored noise white noise

22

Whitening Filter: What is a whitening filter? From the discussion above, the output spectral density of the whitened noise process equals unity, since it represents the normalized white noise by design. But from (18-20) which gives i.e., the whitening filter transfer function satisfies the magnitude relationship in (18-55). To be useful in practice, it is desirable to have the whitening filter to be stable and causal as well. Moreover, at times its inverse transfer function also needs to be implementable so that it needs to be stable as well. How does one obtain such a filter (if any)? [See section 11.1 page 499-502, (and also page 423-424), Text for a discussion on obtaining the whitening filters.]. ( )

nn

S ω

2

1 ( ) ( ) | ( ) | ,

WW

nn

S S G ω ω ω = =

2

1 | ( ) | . ( )

WW

G S ω ω = (18-55)

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( ) G ω Wiener has been exploited in several other problems including prediction, filtering etc.

23

From there, any spectral density that satisfies the finite power constraint and the Paley-Wiener constraint (see Eq. (11-4), Text) can be factorized as where H(s) together with its inverse function 1/H(s) represent two filters that are both analytic in Re s > 0. Thus H(s) and its inverse 1/ H(s) can be chosen to be stable and causal in (18-58). Such a filter is known as the Wiener factor, and since it has all its poles and zeros in the left half plane, it represents a minimum phase factor. In the rational case, if X(t) represents a real process, then is even and hence (18-58) reads ( )

XX

S d ω ω

+∞ −∞

< ∞

∫

(18-56) (18-57)

2

| log ( ) | 1

XX

S d ω ω ω

+∞ −∞

< ∞ +

∫

2

( ) | ( ) | ( ) ( ) |

XX

s j

S H j H s H s

ω

ω ω

=

= = − (18-58) ( )

XX

S ω

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Example 18.3: Consider the spectrum which translates into The poles ( ) and zeros ( ) of this function are shown in Fig 18.12. From there to maintain the symmetry condition in (18-59), we may group together the left half factors as

2 2

( ) ( ) | ( ) ( ) | .

XX XX

s j s j

S S s H s H s

ω ω

ω

= =

≤ = − = −

• (18-59)

2 2 2 4

( 1)( 2) ( ) ( 1)

XX

S ω ω ω ω + − = +

2 2 2 2 4

(1 )(2 ) ( ) . 1

XX

s s S s s − + − = +

• ○

○ ○ ○

× × × ×

○ ○

Fig 18.12

2 s j = 2 s j = − 1 s=+ 1 s = − 1 2 j s − + = 1 2 j s + = 1 2 j s − − = 1 2 j s − = 2 2

1 1 2 2

( 1)( 2 )( 2 ) ( 1)( 2) ( ) 2 1

j j

s s

s s j s j s s H s s s

           

+ −

+ +

+ − + + + = = + +

○

×

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and it represents the Wiener factor for the spectrum Observe that the poles and zeros (if any) on the appear in even multiples in and hence half of them may be paired with H(s) (and the other half with H(– s)) to preserve the factorization condition in (18-58). Notice that H(s) is stable, and so is its inverse. More generally, if H(s) is minimum phase, then ln H(s) is analytic on the right half plane so that gives Thus and since are Hilbert transform pairs, it follows that the phase function in (18-60) is given by the Hilbert axis jω − ( )

XX

S ω

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( )

( ) ( )

j

H A e

ϕ ω

ω ω

−

= (18-60) ln ( ) ln ( ) ( ) ( ) .

j t

H A j b t e dt

ω

ω ω ϕ ω

+∞ −

= − = ∫ ln ( ) ( )cos ( ) ( )sin

t t

A b t t dt b t t dt ω ω ϕ ω ω = =

∫ ∫

cos and sin t t ω ω ( ) ϕ ω ( ) above.

XX

S ω

∆

26

transform of Thus

• Eq. (18-60) may be used to generate the unknown phase function of

a minimum phase factor from its magnitude. For discrete-time processes, the factorization conditions take the form (see (9-203)-(9-205), Text) and In that case where the discrete-time system ( ) <

XX

S d

π π

ω ω

−

∞

∫

(18-63) (18-62) ln ( ) > .

XX

S d

π π

ω ω

−

− ∞

∫

2

( ) | ( ) |

XX

j

S H e ω ω = ( ) ( )

k k

H z h k z

∞ − =

= ∑

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ln ( ). A ω ( ) {ln ( )}. A ϕ ω ω = H (18-61)

27

is analytic together with its inverse in |z| >1. This unique minimum phase function represents the Wiener factor in the discrete-case. Matched Filter in Colored Noise: Returning back to the matched filter problem in colored noise, the design can be completed as shown in Fig 18.13. (Here represents the whitening filter associated with the noise spectral density as in (18-55)-(18-58). Notice that G(s) is the inverse of the Wiener factor L(s) corresponding to the spectrum i.e., The whitened output sg(t) + n(t) in Fig 18.13 is similar h0(t)=sg(t0 – t)

1

( ) ( ) G j L j ω ω

−

=

t t =

( ) ( )

g

s t n t +

( ) ( ) ( ) r t s t w t = +

Whitening Filter Matched Filter

Fig 18.13 ( ) G jω ( )

WW

S ω ( ).

WW

S ω

2

( ) ( ) | | ( ) | ( ).

WW

s j

L s L s L j S

ω

ω ω

=

− = = (18-64)

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28

to (18-38), and from (18-45) the optimum receiver is given by where If we insist on obtaining the receiver transfer function for the

• riginal colored noise problem, we can deduce it easily from Fig 18.14

Notice that Fig 18.14 (a) and (b) are equivalent, and Fig 18.14 (b) is equivalent to Fig 18.13. Hence (see Fig 18.14 (b))

• r

( ) ( )

g

h t s t t = −

1

( ) ( ) ( ) ( ) ( ) ( ).

g g

s t S G j S L j S ω ω ω ω ω

−

↔ = = ( ) H ω ( ) H ω

t t =

≡

L-1(s) L(s) ( ) H ω

t t =

( ) r t ( ) r t

0 (

) H ω

• (a)

(b)

Fig 18.14

0( )

( ) ( ) H L j H ω ω ω =

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29

turns out to be the overall matched filter for the original problem. Once again, transmit signal design can be carried out in this case also.

AM/FM Noise Analysis:

Consider the noisy AM signal and the noisy FM signal where

1 1 * 1 1 *

( ) ( ) ( ) ( ) ( ) ( ){ ( ) ( )}

j t g j t

H L j H L S e L L S e

ω ω

ω ω ω ω ω ω ω ω

− − − − − −

= = = (18-65) ( ) ( )cos( ) ( ), X t m t t n t ω θ = + + (18-66) (18-67) ( ) cos( ( ) ) ( ), X t A t t n t ω ϕ θ = + + + ( ) FM ( ) ( ) PM.

t

c m d t cm t τ τ ϕ   =   

∫

(18-68)

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Here m(t) represents the message signal and a random phase jitter in the received signal. In the case of FM, so that the instantaneous frequency is proportional to the message signal. We will assume that both the message process m(t) and the noise process n(t) are w.s.s with power spectra and respectively. We wish to determine whether the AM and FM signals are w.s.s, and if so their respective power spectral densities. Solution: AM signal: In this case from (18-66), if we assume then so that (see Fig 18.15) ( ) ( ) ( ) t t c m t ω ϕ′ = = θ ( )

mm

S ω ( )

nn

S ω ~ (0,2 ), U θ π 1 ( ) ( )cos ( ) 2

XX mm nn

R R R τ τ ω τ τ = + (18-69) ( ) ( ) ( ) ( ). 2

XX XX XX nn

S S S S ω ω ω ω ω ω − + + = + (18-70)

( )

mm

S ω ω ω ω ω − ( )

XX

S ω ( )

mm

S ω ω −

(a) (b)

Fig 18.15

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Thus AM represents a stationary process under the above conditions. What about FM? FM signal: In this case (suppressing the additive noise component in (18-67)) we obtain since

2 2 2

( / 2, / 2) {cos( ( / 2) ( / 2) ) cos( ( / 2) ( / 2) )} {cos[ ( / 2) ( / 2)] 2 cos[2 ( / 2) ( / 2) 2 ]} [ {cos( ( / 2) ( / 2))}cos 2 {sin( ( / 2) ( / 2))

XX

R t t A E t t t t A E t t t t t A E t t E t t τ τ ω τ ϕ τ θ ω τ ϕ τ θ ω τ ϕ τ ϕ τ ω ϕ τ ϕ τ θ ϕ τ ϕ τ ω τ ϕ τ ϕ τ + − = + + + + × − + − + = + + − − + + + + − + = + − − − + − − }sin ] ω τ (18-71)

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{cos(2 ( / 2) ( / 2) 2 )} {cos(2 ( / 2) ( / 2))} {cos2 } {sin(2 ( / 2) ( / 2))} {sin 2 } 0. E t t t E t t t E E t t t E ω ϕ τ ϕ τ θ ω ϕ τ ϕ τ θ ω ϕ τ ϕ τ θ + + + − + = + + + − − + + + − =

32

Eq (18-71) can be rewritten as where and In general and depend on both t and so that noisy FM is not w.s.s in general, even if the message process m(t) is w.s.s. In the special case when m(t) is a stationary Gaussian process, from (18-68), is also a stationary Gaussian process with autocorrelation function for the FM case. In that case the random variable τ ( , ) a t τ ( , ) b t τ

2

( / 2, / 2) [ ( , )cos ( , )sin ] 2

XX

A R t t a t b t τ τ τ ω τ τ ω τ + − = − (18-72) (18-74) (18-73) ( ) t ϕ

2 2 2

( ) ( ) ( )

mm

d R R c R d

ϕϕ ϕ ϕ

τ τ τ τ

′ ′

− = = (18-75)

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( , ) {cos( ( / 2) ( / 2))} a t E t t τ ϕ τ ϕ τ = + − −

∆

( , ) {sin( ( / 2) ( / 2))} b t E t t τ ϕ τ ϕ τ = + − −

∆

33

where Hence its characteristic function is given by which for gives where we have made use of (18-76) and (18-73)-(18-74). On comparing (18-79) with (18-78) we get and so that the FM autocorrelation function in (18-72) simplifies into (18-76)

2

2( (0) ( )).

Y

R R

ϕϕ ϕϕ

σ τ = − (18-77)

2 2 2

( (0) ( )) /2

{ }

Y

R R j Y

E e e e

ϕϕ ϕϕ τ

ω ω σ ω − − −

= = (18-78) 1 ω = { } {cos } {sin } ( , ) ( , ),

jY

E e E Y jE Y a t jb t τ τ = + = + (18-79)

( (0) ( ))

( , )

R R

a t e

ϕϕ ϕϕ τ

τ

− −

= (18-80) ( , ) b t τ ≡ (18-81)

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2

( / 2) ( / 2) ~ (0, )

Y

Y t t N ϕ τ ϕ τ σ = + − −

∆

34

Notice that for stationary Gaussian message input m(t) (or ), the nonlinear output X(t) is indeed strict sense stationary with autocorrelation function as in (18-82). Narrowband FM: If then (18-82) may be approximated as which is similar to the AM case in (18-69). Hence narrowband FM and ordinary AM have equivalent performance in terms of noise suppression. Wideband FM: This case corresponds to In that case a Taylor series expansion or gives ( ) t ϕ

2 ( (0) ( ))

( ) cos . 2

XX

R R

A R e

ϕϕ ϕϕ τ

τ ω τ

− −

= (18-82) (0) 1, R

ϕϕ

• (

1 , | | 1)

x

e x x

−

− <<

• 2

( ) {(1 (0)) ( )}cos 2

XX

A R R R

ϕϕ ϕϕ

τ τ ω τ = − + (18-83) (0) 1. R

ϕϕ

> ( ) Rϕϕ τ

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35

and substituting this into (18-82) we get so that the power spectrum of FM in this case is given by where Notice that always occupies infinite bandwidth irrespective

• f the actual message bandwidth (Fig 18.16)and this capacity to spread

the message signal across the entire spectral band helps to reduce the noise effect in any band. ( )

XX

S ω

{ }

2 2 2

2 (0)

( ) cos 2

c mm XX

R

A R e

τ

τ ω τ

− +

=

• (18-85)

(18-86)

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{ }

1 2

( ) ( ) ( )

XX

S S S ω ω ω ω ω = − + +

2 2 2

1 ( ) (0) (0) (0) (0) 2 2

mm

c R R R R R

ϕϕ ϕϕ ϕϕ ϕϕ

τ τ τ ′′ = + + = − +

• (18-84)

(18-87)

ω ω ω ( )

XX

S ω

Fig 18.16

2 2

2 / 2 (0)

( ) . 2

mm

c R

A S e ω ω

−

− ~

36

Spectrum Estimation / Extension Problem

Given a finite set of autocorrelations one interesting problem is to extend the given sequence of autocorrelations such that the spectrum corresponding to the overall sequence is nonnegative for all frequencies. i.e., given we need to determine such that Notice that from (14-64), the given sequence satisfies Tn > 0, and at every step of the extension, this nonnegativity condition must be

• satisfied. Thus we must have

Let Then

1

, , , ,

n

r r r

• 1

, , , ,

n

r r r

• 1

2

, ,

n n

r r

+ +

( ) 0.

jk k k

S r e

ω

ω

∞ − =−∞

= ≥

∑

(18-88) 0, 1, 2, .

n k

T k

+ >

=

• 1

.

n

r x

+ =

(18-89)

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so that after some algebra

• r

where

2 2 2 1 1 1 1

| | det

n n n n n n

x T ξ

− + + −

∆ − ∆ − ∆ = = > ∆ (18-90)

2 2 1 1

| | ,

n n n n

r ξ

+ −

  ∆ − <   ∆   (18-91)

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1 1 * * * 1

= , ,

n n n n

x r T T r x r r r

+

• Fig 18.17

1 n

r +

⋅ (18-92)

1 1 1 2 1 1

, [ , , , ] , [ , , , ] .

T T T n n n n n

a T b a r r r b r r r ξ

− − −

= = =

• ∆

∆ ∆

38

• Eq. (18-91) represents the interior of a circle with center and radius

as in Fig 18.17, and geometrically it represents the admissible set of values for rn+1. Repeating this procedure for it follows that the class of extensions that satisfy (18-85) are infinite. It is possible to parameterically represent the class of all admissible spectra. Known as the trigonometric moment problem, extensive literature is available on this topic. [See section 12.4 “Youla’s Parameterization”, pages 562-574, Text for a reasonably complete description and further insight into this topic.].

2 3

, , ,

n n

r r

+ + n

ξ

1

/

n n−

∆ ∆

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